The present application relates to wireless communication apparatuses for receiving radio-frequency (RF) signals modulated by orthogonal frequency division multiplexing (OFDM). In particular, the present invention relates to wireless communication apparatuses for receiving OFDM signals using a direct-conversion architecture in which no intermediate-frequency (IF) stage is used.
More specifically, the present application relates to a wireless communication apparatus for removing a frequency offset using a training sequence added to the header of packets and demodulating OFDM symbols. In particular, the present invention relates to a wireless communication apparatus for accurately estimating a frequency offset in the presence of both time-varying DC offsets and in-phase and quadrature-phase (IQ) imbalance in received OFDM symbols.
Wireless networks have been attracted attention as cable-free systems in place of traditional wired communication systems. IEEE (The Institute of Electrical and Electronics Engineers) 802.11 is a standard commonly used for wireless networks.
For example, when a wireless network is set up in an indoor environment, there occurs a problem in that a receiving apparatus receives a superposition of a direct wave and a plurality of reflected waves and delayed waves, that is, multipath reception occurs. The multipath reception causes delay distortion (or frequency selective fading), resulting in communication errors. The delay distortion causes interference between symbols. In wireless local area network (LAN) standards such as IEEE 802.11a/g, the OFDM modulation scheme, which is one of multicarrier modulation schemes, is adopted (see, for example, IEEE 802.11a, Part 11: Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specifications: High-speed Physical Layer in the 5 GHZ Band; and IEEE 802.11g, Part 11: Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specifications: High-speed Physical Layer in the 2.4 GHZ Band).
TABLEStandardIEEE 802.11aIEEE 802.11gUsed Frequency5.2GHz2.4GHZMaximum Transmission Speed54Mbps54MbpsModulation SchemeOFDMOFDM
An OFDM transmitter converts information transmitted by serial signals into parallel data for each symbol period with a rate lower than an information transmission rate, and assigns the plurality of parallel data streams to subcarriers for modulation of the amplitude and phase for each of the subcarriers. The OFDM transmitter further performs an inverse fast Fourier transform (IFFT) on the plurality of subcarriers to convert the frequency-domain subcarriers into time-domain signals, and transmits the resulting signals. An OFDM receiver performs an operation reverse to the operation of the OFDM transmitter. That is, the OFDM receiver performs a fast Fourier transform (FFT) to convert the time-domain signals into frequency-domain signals for demodulation according to the modulation schemes corresponding to the subcarriers. The OFDM receiver further performs parallel-serial conversion, and reproduces the original information transmitted by the serial signal. The frequencies of the carriers are determined so that the subcarriers are orthogonal to each other over the symbol periods. The subcarriers being orthogonal to each other means that the peak point of the spectrum of a given subcarrier constantly matches the zero points of the spectra of other subcarriers and no crosstalk occurs therebetween. Accordingly, transmission data is transmitted over a plurality of carriers with orthogonal frequencies, and advantages of narrow bandwidths of the carriers, high frequency-use efficiency, and high resistance to frequency selective fading are achieved. An effective OFDM modem can therefore be implemented by using FFT algorithm. The OFDM transmission scheme is used in wireless LAN systems and various other broadband digital communication systems such as terrestrial digital broadcasting systems (see, for example, J. Olsson, “WLAN/WCDMA Dual-Mode Receiver Architecture Design Trade-Offs,” Proc. of IEEE 6th CAS Symp., vol. 2, pp. 725-728, 31 May to 2 Jun., 2004), and fourth-generation mobile communication systems, and power line carrier communication systems.
In the RF front-end of a wireless communication apparatus, during transmission, generally, after an analog baseband signal is up-converted into an RF band signal using a frequency converter (quadrature modulator) and is band-limited using a band-pass filter, the transmission power is amplified using a variable gain amplifier circuit. During reception, the signal received by an antenna is amplified by a low-noise amplifier (LNA), and is then down-converted into a baseband signal using a local frequency fLO. An automatic gain control (AGC) circuit is used to maintain the current of the self signal at an appropriate constant level.
In recent wireless communication apparatuses, a frequency converter that up-converts or down-converts a transmission/reception signal uses a direct-conversion architecture to perform direct frequency conversion using a carrier frequency fc as the local frequency fLO. The direct-conversion architecture does not use an external intermediate frequency (IF) filter (also referred to as an “RF interstage filter”), leading to reduction in size and power consumption and increased integration compared with the superheterodyne structure. Further, in principle, no spurious frequencies are generated, and the direct-conversion architecture is superior in terms of design of the transmitter and receiver. In the direct-conversion receiver architecture, however, there has been pointed out a problem in that due to the equality of the receiving frequency and the local frequency, a direct-current component, or DC offset, is caused at the down-converter output by self-mixing of a local signal (see, for example, Anuj Batra, “03267r1P802-15_TG3a-Multi-band-OFDM-CFP-Presentation.ppt”, pp. 17, Jul. 2003). Self-mixing occurs due to the finite isolation between the local signal and the RF port of the low-noise amplifier or the mixer. The term DC, as used herein, is defined as 0 Hz (zero IF) in a baseband signal in the OFDM modulation scheme.
OFDM communication systems have a problem in that a small error between frequencies of oscillators in a transmitter and a receiver (for example, in a wireless LAN, oscillators with an accuracy of about 20 ppm are used) causes a frequency offset in the receiver. Although subcarriers do not interfere with each other, frequency orthogonality between the subcarriers is not maintained in the presence of frequency offsets, resulting in degradation in the demodulation characteristics, i.e., errors in received data.
In packet-switched wireless communication systems such as an IEEE 802.11 communication system, symbols known to both the transmitter and the receiver, i.e., a training sequence, are placed at the head of each packet. The receiver uses the received training sequence to perform automatic gain control of the low-noise amplifier, DC-offset estimation and removal, frequency-offset estimation and removal, packet detection, and timing detection. If the frequency-offset processing is implemented by an analog circuit, complexity of circuit structure and power consumption are increased. Therefore, the present inventors consider it preferable that the frequency-offset estimation and compensation processing be implemented by digital processing. In response to an observation of frequency offset, the phase of the data is inverted to correct the frequency offset.
The problem of frequency offset will be examined in the context of IEEE 802.11a/g. FIG. 15 shows the preamble structure specified in IEEE 802.11a/g (see, for example, M. Itami, “OFDM Modulation Technique,” Triceps, 2000; and IEEE 802.11a, Part 11: Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specifications: High-speed Physical Layer in the 5 GHZ Band). As shown in FIG. 15, a short preamble period of 8.0 μs and a long preamble period of 8.0 μs are added to the head of a packet. The short preamble period is formed of a short training sequence (STS), in which ten short preamble symbols t1 to t10 are repeatedly transmitted. The long preamble period is formed of a long training sequence (LTS), in which two long preamble symbols T1 to T2 are repeatedly transmitted after a guard interval GI2 of 1.6 μs. One short preamble symbol is formed of 12 subcarriers, and has a length of 0.8 μs, which corresponds to one quarter of an IFFT/FFT period TFFT. One long preamble symbol is formed of 52 subcarriers, and has a length of 3.2 μs, which corresponds to the IFFT/FFT period TFFT. As shown in FIG. 30, an OFDM signal does not include the DC or 0-Hz subcarrier to avoid DC-offset interference.
IEEE 802.11a/g does not specify the use of preambles. In general, a receiver sets the gain of the receiver and corrects a DC offset using four STS symbols of 0.8 μs, and performs frequency-offset estimation and correction, packet detection, and coarse timing detection using the remaining six STS symbols.
The frequency-offset estimation can be derived using the 0.8-microsecond periodicity of the STS according to equation (1) as follows:
                              Δ          ⁢                                          ⁢                      f            ⁡                          (              k              )                                      =                              1                          2              ⁢              π              ⁢                                                          ⁢                              T                STS                                              ⁢                      1            M                    ⁢                                    ∑                              i                =                k                                            k                +                M                -                1                                      ⁢                          arg              ⁢                                                          ⁢                              (                                                      S                    ⁡                                          (                      i                      )                                                        ⁢                                                            S                      ⁡                                              (                                                  i                          -                          16                                                )                                                              *                                                  )                                                                        (        1        )            
where Tsts indicates 0.8 microseconds, S(i) denotes an STS signal sampled at a frequency of 20 MHz, S*(i) denotes the complex conjugate of the STS signal, and M denotes the average number of samples.
Packet detection and coarse timing detection can be performed on the basis of a correlation value normalized by the STS power level, which is given by equation (2) below:
                              CF          ⁡                      (            k            )                          =                              1            N                    ⁢                                    ∑                              i                =                k                                            k                -                M                -                1                                      ⁢                                                            S                  ⁡                                      (                    i                    )                                                  ⁢                                                      S                    ⁡                                          (                                              i                        -                        16                                            )                                                        *                                                                              S                  ⁡                                      (                    i                    )                                                  ⁢                                                      S                    ⁡                                          (                      i                      )                                                        *                                                                                        (        2        )            
where N denotes the average number of samples.
In the packet detection, a threshold level is set for the correlation value given by equation (2) above, and a packet is detected when a correlation value exceeds the threshold level. In the coarse timing detection, the characteristic that the correlation value is changed from the increase to the decrease at the end of the STS is utilized. That is, the currently determined correlation value is compared with the previously determined correlation value to determine a coarse timing.
Accordingly, the receiver performs automatic gain control of the low-noise amplifier, DC-offset estimation and removal, frequency-offset estimation and removal, packet detection, and timing detection using the preamble portion of each packet.
However, the accuracy of frequency-offset estimation, packet detection, and timing detection is sensitive to DC offsets, and there arises a problem in that it is difficult to accurately estimate a frequency offset in the presence of DC offsets. In the above-described direct-conversion architecture in particular, the problem of DC offset caused by self-mixing is serious, and the quality of the received signal may be impaired by both frequency offsets and DC offsets.
For example, in the presence of a DC offset at the I-axis and Q-axis inputs, the correlation value increases even during silent periods. That is, the correlation value is constantly increased by one, and as a result of the continuous increase, the correlation value exceeds the threshold value that packet detection is based on. Therefore, the receiver recognizes that a packet is received even during silent periods, resulting in an operation error.
In the presence of a DC offset at the I-axis and Q-axis inputs, further, even in a portion where received signals are not correlated with each other, the correlation value is not changed from the increase to the decrease due to the influence of the DC offset. Therefore, the coarse-timing-detection characteristics are degraded.
In the presence of a DC offset, further, the accuracy of frequency-offset estimation is lowered, and residual frequency offsets cause further degradation in characteristics of the received signal. A frequency offset that has not been removed causes a phase rotation of the overall subcarriers of the OFDM symbols subsequent to the training sequence, and causes error floor in which packet errors still occur even if the signal-to-noise (SN) ratio is increased. Conversely, when a DC offset is estimated in the presence of a frequency offset, it is difficult to accurately estimate the DC offset. Therefore, it is desirable to address the presence of both DC offsets and frequency offsets.
As described above, it is desirable to perform high-accuracy DC-offset correction in the preceding stage of a frequency-offset correction circuit module within a period of time as short as the first four STS symbols t1 to t4. In general, it is difficult to implement short-time high-accuracy DC-offset correction, and the power consumption and the circuit size are considerably increased.
There have been a method for removing a DC offset using a high-pass filter (HPF), a method for estimating a DC offset and a frequency offset at the same time, a method for estimating a DC offset and a frequency offset in parallel, a method for repeating DC-offset estimation and frequency-offset compensation, and the like.
FIG. 16 schematically shows the structure of a receiver that removes a DC offset using an HPF (see, for example, W. Namgoong and T. H. Meng, “Direct-Conversion RF Receiver Design”, IEEE Trans. on Commun. Vol. 49, No. 3, March 2001). In the receiver shown in FIG. 16, the DC-offset component included in received OFDM symbols is removed using the HPF. Then, signal processing is performed to estimate a frequency offset, and the frequency offset is removed from the OFDM symbols subsequent to the training sequence. In this method, however, the HPF attenuates a near-DC signal in the OFDM symbols, and the demodulation characteristics may be degraded.
One approach to prevent the attenuation of a near-DC signal is to sufficiently reduce the cutoff frequency fc of the HPF relative to the subcarrier spacing (see FIG. 17A). However, if the gain of the low-noise amplifier is changed by automatic gain control, there is a problem in that a time-varying DC offset is generated (see, for example, S. Otaka, T. Yamaji, R. Fujimoto, and H. Tanimoto, “A Low Offset 1.9 GHz Direct Conversion Receiver IC with Spurious Free Dynamic Range of over 67 dB”, IECE Trans. on Fundamentals, vol. E84-A, no. 2, pp. 513-519, February 2001). The HPF with low cutoff frequency fc has low response, and a time-varying DC offset may be transmitted through the HPF.
For example, in the preamble structure shown in FIG. 15, the gain of the low-noise amplifier is changed from high to low around the center of the short preamble period. A DC offset largely varies over time in accordance with the change in gain, and the high-frequency component is included in the DC offset (see part (a) of FIG. 31). Since the HPF with low cutoff frequency fc has low response, the high-frequency component of the time-varying DC offset is transmitted through the HPF, and also passes through a frequency-offset estimator in the subsequent stage. Such a residual DC offset affects fine frequency-offset estimation performed in the long preamble period if it still exists in the subsequent long preamble period (see part (b) of FIG. 31), resulting in low estimation accuracy.
For example, in IEEE 802.11a/g, the preamble period is significantly short, and it is desirable that a residual DC offset be rapidly converged using the HPF. The convergence time is minimized by greatly increasing the cutoff frequency fc of the HPF (see, for example, T. Yuba and Y. Sanada, “Decision Directed Scheme for IQ Imbalance Compensation on OFDCM Direct Conversion Receiver”, IEICE Trans. on Communications, vol. E89-B, no. 1, pp. 184-190, January 2006).
The HPF with an increased cutoff frequency fc has high response to changes in DC offset caused by changing the gain of the low-noise amplifier, and may cut even an effective near-DC signal (see FIG. 17B). Therefore, the OFDM demodulation characteristics may be degraded.
In view of better transient response and fast convergence speed, it is preferable that the cutoff frequency fc of the HPF be high. In the preamble structure shown in FIG. 15, in the STS having a spacing of 1.25 MHz between the DC component and the subcarrier nearest the DC component, the signals of the near-DC subcarriers are not cut even if the cutoff frequency is high. In the subsequent LTS having a spacing of 312.5 kHz between the subcarriers nearest DC, however, the HPF would cut the signals of the near-DC subcarriers, resulting in degradation in the demodulation characteristics.
FIG. 18 schematically shows the structure of a receiver that estimates a DC offset and a frequency offset at the same time (see, for example, G. T. Gil, I. H. Sohn, J. K. Park, and Y. H. Lee, “Joint ML Estimation of Carrier Frequency, Channel, I/Q Mismatch, and DC Offset in Communication Receivers”, IEEE Trans. on Vehi. Tech., Vol. 54, No. 1, January 2005). In the receiver shown in FIG. 18, a DC offset and a frequency offset are simultaneously estimated and compensated using a maximum likelihood estimation method. However, due to the large amount of computation and the long computation time of the maximum likelihood estimation method, it is difficult to implement maximum likelihood estimation in a system with limited offset compensation time. In the preamble structure shown in FIG. 15, it is desirable that DC-offset estimation be completed within about the first four STS symbols t1 to t4, i.e., about 3.2 microseconds.
FIG. 19 schematically shows the structure of a receiver that estimates a DC offset and a frequency offset in parallel (see, for example, C. K. Ho, S. Sun, and P. He, “Low Complexity Frequency Offset Estimation in the Presence of DC Offset”, Proc. of IEEE International Conference on Communications 2003, Vol. 3, pp. 2051-2055, May 2003; and U.S. Patent Application Publication Nos. 2003/0174790 and 2005/0078509). A DC-offset estimator estimates a DC offset by averaging over the overall preambles. A frequency-offset estimator in the subsequent stage calculates the correlation function of the preamble signal, and subtracts the estimated DC offset to estimate an accurate frequency offset from the DC-offset-removed signal. However, if the level of the DC offset is changed by changing the gain of the low-noise amplifier or the like during the reception of preambles, the DC offset may be incorrectly estimated.
FIG. 20 schematically shows the structure of a receiver that repeats DC-offset estimation and frequency-offset compensation (see, for example, U.S. Patent Application Publication Nos. 2005/0020226, 2003/0133518, 2004/0202102, and 2005/0276358). In the receiver shown in FIG. 20, a DC-offset remover removes a DC offset, and then a frequency offset is estimated. After the frequency offset is compensated, a DC offset is further estimated to cancel a residual DC offset. This method takes a long time for convergence of a feedback loop for DC-offset removal, and it is difficult to use the method for short preambles. Further, if a DC offset is varied by changing the gain of the low-noise amplifier or the like, errors occur in the frequency-offset estimation.
As is known, the DC-offset level varies in accordance with a change of the gain set in the low-noise amplifier. The receivers shown in FIGS. 18 to 20 do not take sufficient account of DC offsets varying over time in accordance with changes of the gain in the low-noise amplifier.
OFDM direct-conversion receivers also have a problem of IQ imbalance as well as DC offset caused by self-mixing of a local signal. The direct-conversion architecture does not use IF signals in the digital domain, and IQ quadrature demodulation is not performed in the digital domain but is performed in the analog domain. IQ imbalance is caused by imbalance between in-phase (I) components and quadrature-phase (Q) components. In particular, IQ phase imbalance is caused by non-90-degree phase difference between local signals input to I-channel and Q-channel mixers, and IQ gain imbalance is caused by gain difference between signals in the I channel and Q channel (see, for example, T. Yuba and Y. Sanada, “Decision Directed Scheme for IQ Imbalance Compensation on OFDCM Direct Conversion Receiver”, IEICE Trans. on Communications, vol. E89-B, no. 1, pp. 184-190, January 2006). Like DC offsets, IQ imbalances cause degradation in accuracy of frequency-offset estimation, and also affect the decoding characteristics.
In the OFDM direct-conversion receiver architecture, therefore, the quality of received signals may be impaired due to frequency offset, DC offset, and IQ imbalance. The receivers shown in FIGS. 18 to 20 do not take account of both time-varying DC offset and IQ imbalance.